Orthogonal frequency division multiplexing-code division multiple access system

ABSTRACT

An orthogonal frequency division multiplexing (OFDM)-code division multiple access (CDMA) system is disclosed. The system includes a transmitter and a receiver. At the transmitter, a spreading and subcarrier mapping unit spreads an input data symbol with a complex quadratic sequence code to generate a plurality of chips and maps each chip to one of a plurality of subcarriers. An inverse discrete Fourier transform is performed on the chips mapped to the subcarriers and a cyclic prefix (CP) is inserted to an OFDM frame. A parallel-to-serial converter converts the time-domain data into a serial data stream for transmission. At the receiver, a serial-to-parallel converter converts received data into multiple received data streams and the CP is removed from the received data. A discrete Fourier transform is performed on the received data streams and equalization is performed. A despreader despreads an output of the equalizer to recover the transmitted data.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.11/385,168 filed Mar. 21, 2006, which claims the benefit of U.S.Provisional Application Nos. 60/664,868 filed Mar. 24, 2005, 60/665,442filed Mar. 25, 2005, 60/665,811 filed Mar. 28, 2005 and 60/666,140 filedMar. 29, 2005 the contents of which are hereby incorporated by referenceherein.

FIELD OF THE INVENTION

The present invention is related to a wireless communication system.More particularly, the present invention is related to an orthogonalfrequency division multiplexing (OFDM)-code division multiple access(CDMA) communication system.

BACKGROUND

Wireless communication networks of the future will provide broadbandservices such as wireless Internet access to subscribers. Thosebroadband services require reliable and high-rate communications overtime-dispersive (frequency-selective) channels with limited spectrum andintersymbol interference (ISI) caused by multipath fading. OFDM is oneof the most promising solutions for a number of reasons. OFDM has highspectral efficiency and adaptive coding and modulation can be employedacross subcarriers. Implementation is simplified because the basebandmodulation and demodulation can be performed using simple circuits suchas inverse fast Fourier transform (IFFT) circuits and fast Fouriertransform (FFT) circuits. A simple receiver structure is one of theadvantages of OFDM system, since in some cases only one tap equalizer issufficient to provide excellent robustness in multipath environment. Inother cases, when OFDM is used in conjunction with signal spreadingacross multiple subcarriers, a more advanced equalizer may be required.

OFDM has been adopted by several standards such as Digital AudioBroadcast (DAB), Digital Audio Broadcast Terrestrial (DAB-T), IEEE802.11a/g, IEEE 802.16 and Asymmetric Digital Subscriber Line (ADSL).OFDM is being considered for adoption in standards such as Wideband CodeDivision Multiple Access (WCDMA) for third generation partnershipproject (3GPP) long term evolution, CDMA2000, Fourth Generation (4G)wireless systems, IEEE 802.11n, IEEE 802.16, and IEEE 802.20.

Despite all of the advantages, OFDM has some disadvantages. One majordisadvantage of OFDM is its inherent high peak-to-average power ratio(PAPR). The PAPR of OFDM increases as the number of subcarriersincreases. When high PAPR signals are transmitted through a non-linearpower amplifier, severe signal distortion will occur. Therefore, ahighly linear power amplifier with power backoff is required for OFDM.As a result, the power efficiency with OFDM is low and the battery lifeof a mobile device implementing OFDM is limited.

Techniques for reducing the PAPR of an OFDM system have been studiedextensively. These PAPR reduction techniques include coding, clipping,and filtering. The effectiveness of these methods varies and each hasits own inherent trade-offs in terms of complexity, performance, andspectral efficiency.

SUMMARY

The present invention is related to an OFDM-CDMA system. The systemincludes a transmitter and a receiver. At the transmitter, a spreadingand subcarrier mapping unit spreads an input data symbol with a spreadcomplex quadratic sequence (SCQS) code to generate a plurality of chipsand maps each chip to one of a plurality of subcarriers. An inversediscrete Fourier transform (IDFT) or IFFT unit performs IDFT or IFFT onthe chips mapped to the subcarriers and a cyclic prefix (CP) is insertedto an OFDM frame. A parallel-to-serial (P/S) converter converts thetime-domain data into a serial data stream for transmission. At thereceiver, a serial-to-parallel (S/P) converter converts a received datainto multiple received data streams and the CP is removed from thereceived data. A discrete Fourier transform (DFT) or FFT unit performsDFT or FFT on the received data streams and equalization is performed. Adespreader despreads an output of the equalizer to recover thetransmitted data.

BRIEF DESCRIPTION OF THE DRAWINGS

A more detailed understanding of the invention may be had from thefollowing description of a preferred embodiment, given by way of exampleand to be understood in conjunction with the accompanying drawingwherein:

FIG. 1 is a block diagram of an OFDM-CDMA system in accordance with oneembodiment of the present invention;

FIG. 2 shows the code set of the spread complex quadratic sequence(SCQS) code in accordance with the present invention;

FIG. 3 shows spreading and subcarrier mapping in the system of FIG. 1;

FIG. 4 shows an alternative interpretation of the spreading andsubcarrier mapping in the system of FIG. 1;

FIG. 5 is a block diagram of an OFDM-CDMA system in accordance withanother embodiment of the present invention;

FIG. 6 is a block diagram of an OFDM-CDMA system in accordance with yetanother embodiment of the present invention;

FIG. 7 shows an alternative way for the frequency-domain spreading andsubcarrier mapping in a system of FIG. 6; and

FIG. 8 is a block diagram of an exemplary time-frequency Rake combinerin accordance with the present invention

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention is applicable to wireless communication systemsimplementing OFDM and CDMA such as IEEE 802.11, IEEE 802.16, ThirdGeneration (3G) cellular systems for long term evolution, FourthGeneration (4G) systems, satellite systems, DAB, digital videobroadcasting (DVB), or the like.

The features of the present invention may be incorporated into anintegrated circuit (IC) or be configured in a circuit comprising amultitude of interconnecting components.

The present invention provides an OFDM-CDMA system with an improved PAPRand capacity. The present invention uses a special spreading code, aSCQS code, in spreading input data symbols. The SCQS code comprises twocomponents; a quadratic phase sequence code and an orthogonal (orpseudo-orthogonal) spreading code. Examples of the quadratic phasesequence code, denoted by G, are the Newman phase code (or polyphasecode), a generalized chirp-like sequence (GCL) and a Zadoff-Chusequence. Quadratic phase sequences are called polyphase sequences aswell.

To support a variable spreading factor (VSF), the sequence length of thequadratic phase sequence (or polyphase sequence) is limited as K=2^(k).In some special cases, (such as random access channel or uplink pilots),the sequence length of quadratic phase sequence (or polyphase sequence)can be any arbitrary integer number. Given the number of subcarriersN=2^(n) in the system, consider a sequence length of N as an example.Then, the generic Newman phase code or polyphase code sequence is fixed.The generic Newman phase code sequence is:

$\begin{matrix}{{G_{k} = ^{{- j}\; k^{2}\frac{\pi}{N}}},{k = 0},1,\ldots \mspace{14mu},{N - 1.}} & {{Equation}\mspace{14mu} (1)}\end{matrix}$

More orthogonal Newman phase code sequences are created by shifting thegeneric Newman phase code sequence in phase. The l-th shifted version,(or DFT modulated), of the generic Newman polyphase code sequence is:

$\begin{matrix}{{G_{k}^{(l)} = {^{{- j}\; k^{2}\frac{\pi}{N}} \cdot ^{j\; {kl}\frac{2\pi}{N}}}},{k = 0},1,\ldots \mspace{14mu},{N - 1},{l = 0},1,\ldots \mspace{14mu},{N - 1.}} & {{Equation}\mspace{14mu} (2)}\end{matrix}$

Two Newman phase code sequences with different shifts are orthogonal toeach other.

One example of the orthogonal (or pseudo-orthogonal) spreading code,denoted by H, is Walsh-Hadamard code, which is given by:

$\begin{matrix}{{H_{2} = \begin{bmatrix}1 & 1 \\1 & {- 1}\end{bmatrix}};} & {{Equation}\mspace{14mu} (3)} \\{{H_{2^{m}} = \begin{bmatrix}H_{2^{m - 1}} & H_{2^{m - 1}} \\H_{2^{m - 1}} & {- H_{2^{m - 1}}}\end{bmatrix}},{{{for}\mspace{14mu} m} > 1.}} & {{Equation}\mspace{14mu} (4)}\end{matrix}$

The SCQS code is constructed by combining the quadratic phase code andthe orthogonal (or pseudo-orthogonal) spreading code. For a specificspreading factor 2^(m), the SCQS code has 2^(m) chips. The genericquadratic phase sequence code part of the SCQS code has 2^(m) chips,which is:

{G_(i), G₂ _(n−m) _(+i), . . . , G₂ _(n−m) _(·k+i), . . . , G₂ _(m−1)_(+i)};  Equation (5)

where k=0, 1, . . . , 2^(m)−1, i=0, 1, . . . , 2^(n−m)−1.

The l-th shifted version of the quadratic phase sequence code part has2^(m) chips, which is:

{G_(i) ^((l)), G₂ _(n−m) _(+i) ^((l)), . . . , G₂ _(n−m) _(·k+i) ^((l)),. . . , G₂ _(m−1) _(+i)};  Equation (6)

where l=0, 1, . . . , N−1, k=0, 1, . . . , 2^(m)−1, i=0, 1, . . . ,2^(n−m)−1.

For a specific SCQS code with spreading factor 2^(m), the orthogonal (orpseudo-orthogonal) spreading code part of the SCQS code is given by oneof the codes in the orthogonal (or pseudo-orthogonal) spreading code setof spreading factor 2^(m). For example, the h-th code is denoted by H₂_(m) (h,:).

The k-th chip of the SCQS code c_(i) is constructed as a product of thek-th quadratic phase sequence code of the l-th shifted version of thegeneric quadratic phase sequence code and the k-th chip of the h-thorthogonal (or pseudo-orthogonal) spreading code with the size ofN=2^(m).

c _(i) ^(k) =G _(k) ^((l)) ·H ₂ _(m) (h,k),k=0, 1, . . . ,2^(m)−1.  Equation (7)

The code set size of the SCQS code is determined by the code setdimensions of the orthogonal (or pseudo-orthogonal) spreading code partand the quadratic phase sequence code part. The code set dimension ofthe quadratic phase sequence code is fixed regardless of the spreadingfactor and is determined by the number of different shifts, which is thenumber of subcarriers in the system, 2^(n). The code set dimension ofthe orthogonal (or pseudo-orthogonal) spreading code depends on thespreading factor. For example, in the case of a Walsh-Hadamard code, thedimension equals to the spreading factor 2^(m)(0≦m≦n).

Different users are assigned different SCQS codes. In order for areceiver to distinguish between different users, the SCQS codes used bytwo users may be different in the quadratic phase sequence code part,the orthogonal (or pseudo-orthogonal) spreading code part, or both. Thecode set of the SCQS code is shown in FIG. 2.

Without multipath, different SCQS codes are orthogonal as long as theirquadratic phase sequence code parts are different; or an orthogonalspreading code is used. Different SCQS codes are pseudo-orthogonal onlywhen their quadratic phase sequence code parts are the same and apseudo-orthogonal spreading code is used. In both cases, the multipleaccess interference (MAI) between different codes is either zero or verysmall.

Under the multipath fading environment, codes assigned to differentusers should be such that the difference in the shift of quadratic phasesequence code part should be as large as possible. Codes assigned todifferent users should be such that if the difference in the shift ofthe quadratic phase sequence code part of two codes is not less than themaximum delay spread of the multipath channel, there is no MAI betweenthe two codes. Therefore, the corresponding orthogonal (orpseudo-orthogonal) spreading code part can be assigned to be the same.Optionally, the difference in the shift of the quadratic phase sequencecode part may be limited to be at most the maximum delay spread of themultipath channel. This will create more codes with perfect MAIimmunity. This is achievable as long as the number of users in thesystem is no more than N/L, where N is the number of subcarriers and Lis the multipath channel maximum delay spread.

If the difference in the shift of the quadratic phase sequence code partof two codes is less than the maximum delay spread of the multipathchannel, the corresponding orthogonal (or pseudo-orthogonal) spreadingcode part should be different in order to reduce the MAI that cannot becancelled by the difference in the shift of the quadratic phase sequencecode part.

In this way, the MAI can be reduced as compared to the conventional CDMAsystem since the correlation between orthogonal codes is further reducedby the correlation of two quadratic phase sequence codes. For aninterference-limited system (such as CDMA), reduced MAI impliesincreased system capacity.

An OFDM-CDMA system of the present invention comprises a transmitter anda receiver. The transmitter comprises a spreading and subcarrier mappingportion and an OFDM portion. The spreading and subcarrier mappingportion performs spreading of input data symbols into a plurality ofchips and mapping of the chips to one of a plurality of subcarriers. TheOFDM portion performs conventional OFDM operation. The spreading may beperformed in the frequency-domain, in the time-domain or both, whichwill be explained in detail hereinafter.

FIG. 1 is a block diagram of an OFDM-CDMA system 100 in accordance witha first embodiment of the present invention. The system 100 comprises atransmitter 110 and a receiver 150. The transmitter 110 comprises aspreader 112, a serial-to-parallel (S/P) converter 114, a subcarriermapping unit 116, an IDFT unit 118, a cyclic prefix (CP) insertion unit120, a parallel-to-serial (P/S) converter 122 and an optional mixer 124.The spreader 112 spreads input data symbols 101 in frequency-domainusing the SCQS code 111. The procedure of spreading and subcarriermapping is shown in FIG. 3. The spreading factor used by the SCQS codec_(i) is 2^(m) (0≦m≦n). One user can use all of 2^(n) subcarriers in thesystem. Therefore, the number of data symbols that can be transmitted byone user in one OFDM frame is 2^(n−m). Each data symbol d(i) 101 isspread by the spreading code c_(i) 111 into 2^(m) chips 113. The 2^(m)chips 113 are then converted into 2^(m) parallel chips 115 by the S/Pconverter 114 and each chip is mapped to one of the subcarriers 117 bythe subcarrier mapping unit 116 in an equal-distance. The distancebetween each subcarrier used by chips of the same data symbol is 2^(n−m)subcarriers. Chips of different data symbols are mapped to subcarriersin the system sequentially such that the chips of data symbol d(i) aremapped to subcarriers 2^(n−m)·k+i, (k=0, 1, . . . , 2^(m)−1, i=0, 1, . .. , 2^(n−m)−1).

FIG. 4 shows an alternative embodiment for spreading and subcarriermapping. Instead of the spreader 112, a repeater 402 is used to repeateach data symbol d(i) 2^(m) times at the chip rate. The repeated datasymbols 404 are converted into 2^(m) parallel symbols 407 by the S/Pconverter 406 and each symbol is mapped to one of the 2^(m) subcarriersof equal distance by the subcarrier mapping and weighting unit 408sequentially. The distance between each subcarrier is 2^(n−m)subcarriers. Chips of different data symbols are mapped to subcarriersin the system sequentially such that the chips of data symbol d(i) aremapped to subcarriers 2^(n−m)·k+i, (k=0, 1, . . . , 2^(m)−1, i=0, 1, . .. , 2^(n−m)−1). A symbol mapped on each subcarrier 2^(n−m)·k+i isweighted by an SCQS code such that a symbol on subcarrier 2^(n−m)·k+i ismultiplied with the k-th chip of the SCQS code, denoted by c_(i) ^(k).

Referring back to FIG. 1, chips 117 mapped on subcarriers are fed intothe IDFT unit 118 to be converted into time-domain data 119. A cyclicprefix (CP) is then added by the CP insertion unit 120 to the end ofeach OFDM frame. The time-domain data with CP 121 is then converted bythe P/S converter 122 into a serial data 123 and transmitted over thewireless channel. It should be noted that the IDFT operation may bereplaced by IFFT or other similar operations and the CP insertion may beperformed after the IDFT output is converted into a serial data streamby the P/S converter 122 and the CP removal may be performed before thereceived signals are converted to a parallel data stream by the S/Pconverter 154.

Due to the structure of spread data, the IDFT operation can besimplified. The output 119 of the IDFT unit 118 comprises data symbolsshifted by a particular phase. The phase is a function of correspondinginput data subcarrier and data symbol indexes. Therefore, the IDFToperation can be replaced by the computation of the phase shift, whichrequires less computation.

For example, assume n/2<m≦n and the orthogonal (or pseudo-orthogonal)spreading code part of the SCQS code are {1, 1, . . . , 1}. Then, theh-th output of the IDFT unit 118 is given as follows:

$\begin{matrix}{{{{IDFT}(h)} = {{d(i)} \cdot ^{j\frac{{({{p \cdot 2^{n - m}} + i})}^{2} - 2^{n - 2}}{2^{n}}\pi}}};} & {{Equation}\mspace{14mu} (8)}\end{matrix}$

where the value of h satisfies the following condition:

h=2^(n−m) ·p+i,p=0, 1, . . . , 2^(m)−1,i=0, 1, . . . , 2^(n−m)−1.

It is optional to perform the masking operation at the transmitter 110and the corresponding demasking operation at the receiver 150. Thepurpose of masking is to reduce the inter-cell MAI. At the transmitter110, the mixer 124 multiplies the data 123 with a masking code 125before transmission. The corresponding demasking operation is performedat the receiver 150. A mixer 152 multiplies the received signals 128with the conjugate 151 of the masking code 125 to generate a demaskeddata stream 153.

Referring to FIG. 1, the receiver 150 comprises an optional mixer 152,an S/P converter 154, a CP removing unit 156, a DFT unit 158, anequalizer 160 and a despreader (including multipliers 162, a summer 164and a normalizer 166). The time-domain received data 128 are convertedinto parallel data stream 155 by the S/P converter 154 and the CP isremoved by the CP removing unit 156. The performance of these operationsmay be switched as explained hereinabove. The output 157 from the CPremoving unit 156 is then fed into the DFT unit 158 to be converted intofrequency-domain data 159. Equalization on the frequency-domain data 159is performed by the equalizer 160. As in a conventional OFDM system, asimple one-tap equalizer may be used for the frequency-domain data 159at each subcarrier. It should be noted that the DFT operation may bereplaced by an FFT operation or other similar operation.

Due to the structure of spread data, the DFT operation can also besimplified. The outputs 159 of the DFT unit are data symbols shifted bya particular phase. The phase is a function of corresponding input datasubcarrier and data symbol indexes. Therefore, the DFT operation can bereplaced by the computation of the phase shift, which requires lesscomputation. The way it is done is similar, but opposite, to the IDFToperation at the transmitter side.

The equalized data is despread at the frequency-domain. The output 161at each subcarrier after equalization is multiplied by the multipliers162 with the conjugate 168 of the corresponding chip of the SCQS code,c_(i) ^(k), k=0, 1, . . . , 2^(m)−1, used at the transmitter 110. Then,the multiplication outputs 163 at all subcarriers are summed up by thesummer 164 and the summed output 165 is normalized by the normalizer 166by the spreading factor of the SCQS code to recover the data 167.

The receiver 150 may further include a block linear equalizer or a jointdetector (not shown) for processing the output of the despreader. Anytype of block linear equalizer or joint detector may be used. Oneconventional configuration for a block linear equalizer or a jointdetector is the minimum mean square error (MMSE) block linear equalizer.In this case, a channel matrix H is established and computed forsubcarriers, and equalization is performed using the established channelmatrix such that:

{right arrow over (d)}=(H ^(H) H+σ ² I)⁻¹ H ^(H) {right arrow over(r)};  Equation (9)

where H is the channel matrix, {right arrow over (r)} is the receivedsignal in subcarriers and {right arrow over (d)} is the equalized datavector in subcarriers.

For uplink operation, it is preferred to keep a constant envelope afterIDFT operation, which allows use of an efficient and inexpensive poweramplifier. In order to keep a constant envelope, the followingconditions for a system with N=2^(n) subcarriers have to be met. First,the spreading factor 2^(m) is limited by └n/2┘≦m≦n, wherein the term └a┘means the smallest integer larger than a. Second, for spreading factor2^(m), only a fraction of orthogonal codes are used to combine with thequadratic phase sequence codes to generate the SCQS codes that yieldconstant envelope. For example, in the case of Newman phase code andHadamard code, only the first 2^(┌m/2┐) codes of the Hadamard code sets(of size 2^(m)) are used to combine with the Newman phase sequence codeto generate the SCQS codes. The term ┌b┐ means the largest integersmaller than b.

As stated above, as long as the number of users in the system is no morethan N/L, there is no MAI and there is no need to implement multi-userdetection (MUD). When the number of users in the system is more thanN/L, then there will be MAI and MUD may be implemented. The MAI will bemore benign than conventional CDMA system with the same number of users.

Suppose that there are M users in the system. The number of users forMUD in the conventional CDMA system will be M. However, the number ofusers for MUD in the OFDM-CDMA system in accordance with the presentinvention will be ┌M/L┐, which is reduced by a scale of L as compared toa conventional CDMA system. In this way, the complexity of MUD operationis much lower than the MUD in a prior art CDMA system. It is alsopossible to use multiple antennas at the transmitter and/or receiver.

FIG. 5 is a block diagram of an OFDM-CDMA system 500, (multi-carrierdirect sequence (MC-DS) CDMA system), in accordance with a secondembodiment of the present invention. The system 500 comprises atransmitter 510 and a receiver 550. The transmitter 510 comprises an S/Pconverter 512, a plurality of multipliers 514, a sub-carrier mappingunit 516, an IDFT unit 518, a P/S converter 520, a CP insertion unit 522and an optional mixer 524. If there are N=2^(n) subcarriers in thesystem 500, the N consecutive data symbols 501 of the user i areconverted from serial to N parallel symbols 513 by the S/P converter512. The j-th data symbol of the N parallel data symbols 513 of the useri is denoted by d^(j)(i), where j=0, 1, . . . , N−1. The SCQS code usedby the user i is denoted by c_(i). Each of the N parallel data symbols513 is spread in time-domain using the SCQS code c_(i) 511. Thespreading factor of the SCQS code c_(i) is 2^(m) (0≦m≦n), therefore eachdata symbol 513 is spread by the SCQS code c_(i) 511 into 2^(m) chips515.

At each chip duration, one chip of each of the N data symbols d^(j)(i)is transmitted on its corresponding subcarrier j. One user can use allof 2^(n) subcarriers in the system. Therefore, the number of datasymbols that can be transmitted by one user in one OFDM frame is 2^(n).

The chips 515 are mapped to subcarriers by the subcarrier mapping unit516. Chips 517 on subcarriers are fed into the IDFT unit 518, andconverted into time-domain data 519. The time-domain data 519 areconverted from parallel into serial data 521 by the P/S converter 520,and a CP is added to the end of each frame by the CP insertion unit 522.The data with CP 523 is transmitted over the wireless channel. It isequivalent to perform the conventional DS-CDMA operation on eachsubcarrier independently using the SCQS code, and DS-CDMA signals onsubcarriers are transmitted in parallel using OFDM structure.

The receiver 550 comprises a CP removal unit 554, an S/P converter 556,a DFT unit 558, an equalizer 560, a plurality of rake combiners 562, anda P/S converter 564. First, the CP is removed by the CP removing unit554 from the received data 528 via the wireless channel. The data 555 isthen converted from serial to parallel data 557 by the S/P converter556. The parallel data 557 is fed into the DFT unit 558, and convertedto frequency-domain data 559. Then, equalization is applied to thefrequency-domain data 559 by the equalizer 560. As in a conventionalOFDM system, a simple one-tap equalizer may be used at each subcarrier.

Data 561 on each subcarrier after equalization is recovered by Rakecombiners 562, (which include despreaders), in the time-domain. Then,parallel data symbols 563 yielded by each Rake combiners 562 areparallel-to-serial converted by the P/S converter 564 to recover thetransmitted data.

As in the first embodiment of FIG. 1, it is optional to perform amasking operation at the transmitter 510 and the corresponding demaskingoperation at the receiver 550 to reduce the intercell MAI. The mixer 524multiplies output 523 from the CP insertion unit 522 with a masking code525 before transmission. The mixer 552 of the receiver 550 multipliesthe received signals 528 with the conjugate 551 of the masking code usedat the transmitter 510.

FIG. 6 is a block diagram of an OFDM-CDMA system 600 in accordance witha third embodiment of the present invention. The system 600 comprises atransmitter 610 and a receiver 650. The transmitter 610 includes an S/Pconverter 612, a plurality of multipliers 614, a plurality of repeaters616, a plurality of S/P converters 618, a subcarrier mapping andweighting unit 620, an IDFT unit 622, a P/S converter 624, a CPinsertion unit 626 and an optional mixer 628. In accordance with thethird embodiment, the input data symbol is spread twice, once at thetime-domain and the other at the frequency-domain. Assume the totalnumber of subcarriers is 2^(n) and the spreading factors used in thetime-domain and frequency-domain spreading are 2^(p) and 2^(m),respectively. The N_(T) consecutive data symbols 601 of the user i areconverted from serial to parallel N_(T) symbols 613 by the S/P converter612. The value of N_(T) equals to 2^(n−m). The j-th data symbol of theN_(T) parallel data symbols 613 of the user i is denoted by d^(j)(i),where j=0, 1, . . . , N−1. The time-domain spreading code 611 used bythe user i is denoted by H₂ _(p) (i,:). Each of the N_(T) parallel datasymbols 613 is then spread in the time-domain by the multipliers 614 bymultiplying the symbols 613 with the time-domain spreading code H₂ _(p)(i,:) 611. The spreading factor of the time domain spreading code H₂_(p) (i,:) is 2^(p) as defined in Equations (3) and (4). Each datasymbol 613 is spread into 2^(p) chips and N_(T) parallel 2^(p) chipstreams 615 are generated.

After the time-domain spreading, a frequency-domain spreading isperformed. Given the user i, for each chip stream j, (corresponding tothe j-th data symbols of the N_(T) data symbols), at each chip duration,each chip of the N_(T) chip streams is repeated 2^(m) times by therepeater 616 and the 2^(m) repeated chips are converted into parallel2^(m) chips 619 by the S/P converter 618. The 2^(m) chips are thenmapped to 2^(m) equal-distant subcarriers sequentially by the subcarriermapping and weighting unit 620. The distance between each subcarrier is2^(n−m) subcarriers. Subcarrier mapping is performed sequentially suchthat the repeated chips from the j-th chip stream are mapped tosubcarriers 2^(n−m)·k+j, (k=0, 1, . . . , 2^(m)−1, j=0, 1, . . . ,2^(n−m)−1). Before the IDFT operation, a chip on each subcarrier2^(n−m)·k+j is weighted by the k-th chip of the SCQS code c_(i), denotedby c_(i) ^(k).

One user can use all of 2^(n) subcarriers in the system. Therefore, thenumber of data symbols that can be transmitted by one user in one OFDMframe is 2^(n−m).

FIG. 7 shows an alternative way for the frequency-domain spreading andsubcarrier mapping in a system of FIG. 6. Instead of repeating the chips2^(m) times, the chips 615 are directly spread by the frequency-domainspreading code c_(i) ^(k). Given the user i, for each chip stream j,(corresponding to the j-th data symbols of the N_(T) data symbols), ateach chip duration, each of the chips 615 is spread by the SCQS codec_(i) ^(k) 703 into 2^(m) chips 704 by the multipliers 702 and thefrequency-domain spread chips 704 are converted into 2^(m) parallelchips 707 by the S/P converter 706. These parallel chips 707 are thenmapped to 2^(m) equal-distant subcarriers 709 by the subcarrier mappingunit 708 sequentially, as explained hereinabove. The distance betweeneach subcarrier is 2^(n−m) subcarriers. Subcarrier mapping is performedsequentially such that the repeated chips from the j-th chip stream aremapped to subcarriers 2^(n−m)·k+j, (k=0, 1, . . . , 2^(m)−1, j=0, 1, . .. , 2^(n−m)−1).

Referring again to FIG. 6, chips 621 mapped on subcarriers are fed intothe IDFT unit 622, and converted into time-domain data 623. Thetime-domain data 623 is converted from parallel data into serial data625 by the P/S converter 624, and a CP is added to the end of each frameof the data 625 by the CP insertion unit 626. The data with the CP 627is transmitted over the wireless channel.

The receiver 650 includes an optional mixer 652, a CP removal unit 654,an S/P converter 656, a DFT unit 658, an equalizer 660, a plurality oftime-frequency Rake combiners 662 and a P/S converter 664. At thereceiver 650 side, the CP is removed by the CP removal unit 654 from thereceived data 632 via the wireless channel. The data 655 is thenconverted from serial to parallel data 657 by the S/P converter 656. Theparallel data 657 is fed into the DFT unit 658, and converted tofrequency-domain data 659. Then, equalization is applied to thefrequency-domain data 659 by the equalizer 660. As in a conventionalOFDM system, a simple one-tap equalizer may be used at each subcarrier.

After equalization, data 661 on each subcarrier is recovered bytime-frequency Rake combiners 662, which will be explained in detailhereinafter. Parallel data symbols 663 yielded by each of thetime-frequency Rake combiners 662 are then parallel-to-serial convertedby the P/S converter 664 to recover the transmitted data.

A time-frequency Rake combiner 662 is a Rake combiner that performsprocessing at both the time and frequency domains in order to recoverthe data that is spread in both the time and frequency domains at thetransmitter. FIG. 8 shows exemplary time-frequency Rake combiners 662.It should be noted that the time-frequency Rake combiners 662 may beimplemented in many different ways and the configuration shown in FIG. 8is provided as an example, not as a limitation, to those of ordinaryskill in the art.

Each time-frequency Rake combiner 662 comprises a subcarrier groupingunit 802, a despreader 804 and a Rake combiner 806. For each data symbolj (j=0, 1, . . . , 2^(n−m)−1) of N_(T) consecutive data symbols, thesubcarrier grouping unit 802 collects the following chips on subcarriers661 2^(n−m)·k+j, (k=0, 1, . . . , 2^(m)−1), totaling 2^(m) chips. Then,the despreader 804 performs frequency-domain despreading to the chips onthe 2^(m) subcarriers. The despreader 804 includes a plurality ofmultipliers 812 for multiplying conjugate 813 of the SCQS codes to thecollected chips 811, a summer 815 for summing the multiplication outputs814, and a normalizer 817 for normalizing the summed output 816. Afterthe frequency-domain despreading, chips on 2^(n) subcarriers becomechips on N_(T) parallel chip streams 818. To recover the j-th datasymbol of the user i, time-domain Rake combining is performed by theRake combiner 806 on the corresponding chip stream 818.

Referring again to FIG. 6, it is optional to perform a masking operationat the transmitter 610 and the corresponding demasking operation at thereceiver 650 to reduce the intercell MAI. The mixer 628 multipliesoutput 627 from the CP insertion unit 626 with a masking code 630 beforetransmission. The mixer 652 of the receiver 650 multiplies the receivedsignals 632 with the conjugate 651 of the masking code used at thetransmitter 610.

For all the embodiments described hereinbefore, a predetermined datavector {d(i)}, (i.e., pre-known signals), may be transmitted. In thisway, the uplink transmitted signals can be used as a preamble for RandomAccess Channel (RACH) or uplink pilot signals. For example, apredetermined data vector {d(i)} of all 1s, {1, 1, . . . , 1}, may betransmitted.

Although the features and elements of the present invention aredescribed in the preferred embodiments in particular combinations, eachfeature or element can be used alone without the other features andelements of the preferred embodiments or in various combinations with orwithout other features and elements of the present invention.

1. A transmitting device, comprising: at least one circuit configured tocombine a data symbol with a first sequence and to spread a combinedresult using an orthogonal sequence, wherein the first sequence isderived by shifting a second sequence having a plurality of phases,wherein the at least one circuit is further configured to map a spreadresult to a plurality of subcarriers, and wherein the at least onecircuit is further configured to process the mapped plurality ofsubcarriers and transmit a result of the processing as a radio frequencysignal.
 2. The transmitting device of claim 1, wherein the secondsequence is a four phase sequence having constant amplitude.
 3. Thetransmitting device of claim 1, wherein the radio frequency signal isfurther derived from a third sequence.
 4. The transmitting device ofclaim 3, wherein the third sequence reduces intercellular interference.5. The transmitting device of claim 3, wherein the third sequence is amasking code.
 6. The transmitting device of claim 1, wherein theorthogonal sequence is assigned to the transmitting device.
 7. Thetransmitting device of claim 1, wherein the at least one circuit isfurther configured to use orthogonal sequences of different lengths forspreading.
 8. The transmitting device of claim 1, wherein the mappedplurality of subcarriers are transmitted as an orthogonal frequencydivision multiplexed (OFDM) based signal.
 9. A method, comprising:combining, by a transmitting device, a data symbol with a firstsequence, wherein the first sequence is derived by shifting a secondsequence having a plurality of phases; spreading, by the transmittingdevice, a combined result using an orthogonal sequence; mapping, by thetransmitting device, a spread result to a plurality of subcarriers;processing, by the transmitting device, the mapped plurality ofsubcarriers; and transmitting a result of the processing as a radiofrequency signal.
 10. The method of claim 9, wherein the second sequenceis a four phase sequence having constant amplitude.
 11. The method ofclaim 9, wherein the radio frequency signal is further derived from athird sequence.
 12. The method of claim 11, wherein the third sequencereduces intercellular interference.
 13. The method of claim 11, whereinthe third sequence is a masking code.
 14. The method of claim 9, whereinthe orthogonal sequence is assigned to the transmitting device.
 15. Themethod of claim 9, wherein the mapped plurality of subcarriers aretransmitted as an orthogonal frequency division multiplexed (OFDM) basedsignal.
 16. A receiving device, comprising: at least one circuitconfigured to receive a radio frequency signal and determine a datasymbol from the radio frequency signal, wherein the radio frequencysignal is a result of combining the data symbol with a first sequence,spreading a combined result using an orthogonal sequence, mapping aspread result to a plurality of subcarriers, and processing the mappedplurality of subcarriers, the first sequence being derived by shifting asecond sequence having a plurality of phases.
 17. The receiving deviceof claim 16, wherein the second sequence is a four phase sequence havingconstant amplitude.
 18. The receiving device of claim 16, wherein theradio frequency signal is further derived from a third sequence.
 19. Thereceiving device of claim 18, wherein the third sequence reducesintercellular interference.
 20. The receiving device of claim 18,wherein the third sequence is a masking code.
 21. The receiving deviceof claim 16, wherein the radio frequency signal is received as anorthogonal frequency division multiplexed (OFDM) based signal.